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| NCP1442, NCP1443, NCP1444, NCP1445 4.0 A 280 kHz/560 kHz Boost Regulators The NCP1442/3/4/5 products are 280 kHz/560 kHz switching regulators with a high efficiency, 4.0 A integrated switch. These parts operate over a wide input voltage range, from 2.7 V to 30 V. The flexibility of the design allows the chips to operate in most power supply configurations, including boost, flyback, forward, inverting, and SEPIC. The ICs utilize current mode architecture, which allows excellent load and line regulation, as well as a practical means for limiting current. Combining high-frequency operation with a highly integrated regulator circuit results in an extremely compact power supply solution. The circuit design includes provisions for features such as frequency synchronization, shutdown, and feedback controls for either positive or negative voltage regulation. 1 Part Number NCP1442 NCP1443 NCP1444 NCP1445 Frequency 280 kHz 280 kHz 560 kHz 560 kHz Feedback Voltage Polarity Positive Negative Positive Negative NC P144xF AWLYWW NC P144xT AWLYWW 7 http://onsemi.com 1 7 PowerFLEX] 7-PIN F SUFFIX CASE 936J 7 LEAD, TO-220 T SUFFIX CASE 821P PIN CONNECTIONS AND MARKING DIAGRAMS Features * * * * * * * * * * * * * * * * Pb-Free Packages are Available* Integrated Power Switch: 4.0 A Guaranteed Wide Input Range: 2.7 V to 30 V High Frequency Allows for Small Components Minimum External Components Easy External Synchronization Built-in Overcurrent Protection Frequency Foldback Reduces Component Stress During an Overcurrent Condition Thermal Shutdown with Hysteresis Regulates Either Positive or Negative Output Voltages Shut Down Current: 50 mA Maximum 1 7 PowerFLEX 7-PIN 1 7 7 LEAD, TO-220 NCP1442/4 Pin 1. VC 2. FB 3. TEST 4. GND 5. VSW 6. SS 7. VCC x A WL Y WW NCP1443/5 1. VC 2. TEST 3. NFB 4. GND 5. VSW 6. SS 7. VCC Applications Boost Converter Inverting Converter Distributed Power Portable Computers Battery Powered Systems = Device Number 2, 3, 4, or 5 = Assembly Location = Wafer Lot = Year = Work Week ORDERING INFORMATION See detailed ordering and shipping information in the package dimensions section on page 18 of this data sheet. *For additional information on our Pb-Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. (c) Semiconductor Components Industries, LLC, 2004 1 October, 2004 - Rev. 7 Publication Order Number: NCP1442/D NCP1442, NCP1443, NCP1444, NCP1445 3.3 V + + + 10 mH 33 mF 33 mF 33 mF 1 VSW MBRS320T3 7 NCP1442/4 VC NC SS VCC VSW 5 VOUT/1.5 A 5 22 k 2 3 FB 4 220 pF 0.01 mF SS 6 GND 7.5 k + + + 33 mF 33 mF 33 mF GND 5.1 k GND Figure 1. Application Diagram - NCP1442/4, 3.3 V to 5.0 V/1.5 A Boost Converter MAXIMUM RATINGS Rating Thermal Resistance Junction-to-Air, TO220-7 Version In Air (Socketed) Thermal Resistance Junction-to-Air, TO220-7 Version On Cold Plate (25C) Thermal Resistance Junction-to-Air, PowerFLEX on 2.1 sq. in. 1 oz. Junction Temperature Range, TJ Storage Temperature Range, TSTORAGE Lead Temperature Soldering: Reflow (Note 1) ESD, Human Body Model Value 66.7 1.45 53.8 0 to +150 -65 to +150 230 Peak 2.0 Unit C/W C/W C C C kV Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied, damage may occur and reliability may be affected. 1. 60 second maximum above 183C. MAXIMUM RATINGS Pin Name IC Power Input Shutdown/Sync Loop Compensation Voltage Feedback Input Pin Symbol VCC SS VC FB (NCP1442/4 only) NFB (NCP1443/5 only) Test GND VSW VMAX 30 V 30 V 6.0 V 10 V VMIN -0.3 V -0.3 V -0.3 V -0.3 V ISOURCE N/A 1.0 mA 10 mA 1.0 mA ISINK 200 mA 1.0 mA 10 mA 1.0 mA Negative Feedback Input (Transient, 10 ms) Test Pin Ground Switch Input 10 V -10 V 1.0 mA 1.0 mA 6.0 V 0.3 V 40 V -0.3 V -0.3 V -0.3 V 1.0 mA 9.0 A 10 mA 1.0 mA 10 mA 9.0 A http://onsemi.com 2 NCP1442, NCP1443, NCP1444, NCP1445 ELECTRICAL CHARACTERISTICS (2.7 V < VCC < 30 V; 0C < TA < 85C; 0C < TJ < 125C; For all NCP1442/3/4/5 specifications unless otherwise stated.) (See Note 2) Characteristic Positive and Negative Error Amplifiers FB Reference Voltage (NCP1442/4 only) NFB Reference Voltage (NCP1443/5 only) FB Input Current (NCP1442/4 only) NFB Input Current (NCP1443/5 only) FB Reference Voltage Line Regulation (NCP1442/4 only) NFB Reference Voltage Line Regulation (NCP1443/5 only) Positive Error Amp Transconductance Negative Error Amp Transconductance Positive Error Amp Gain Negative Error Amp Gain VC Source Current VC Sink Current VC High Clamp Voltage VC Low Clamp Voltage VC Threshold Oscillator Base Operating Frequency Reduced Operating Frequency Maximum Duty Cycle Base Operating Frequency Reduced Operating Frequency Maximum Duty Cycle FB Frequency Shift Threshold NFB Frequency Shift Threshold Sync/Shutdown Sync Range Sync Range Sync Pulse Transition Threshold SS Bias Current NCP1442/3 NCP1444/5 Rise time = 20 ns SS = 0 V SS = 3.0 V - 2.7 V VCC 12 V 12 V < VCC 30 V - - - -10 - 0.50 12 12 500 1000 2.5 -1.0 0.2 0.85 100 40 - - - - 4.0 1.20 500 400 kHz kHz V mA mA V ms ms NCP1442/3, FB = 1.0 V or NFB = -1.9 V NCP1442/3, FB = 0 V or NFB = 0 V NCP1442/3 NCP1444/5, FB = 1.0 V or NFB = -1.9 V NCP1444/5, FB = 0 V or NFB = 0 V NCP1444/5 Frequency drops to reduced operating frequency Frequency drops to reduced operating frequency 240 30 90 480 60 82 0.36 -0.80 280 68 96 560 120 92 0.40 -0.68 320 120 - 640 160 - 0.44 -0.50 kHz kHz % kHz kHz % V V VC tied to FB; measure at FB VC = 1.25 V FB = VREF NFB = NVREF VC = FB 1.246 -2.60 -1.0 -16 -0.03 1.276 -2.475 0.1 -10 0.01 1.300 -2.40 1.0 -5.0 0.03 V V mA mA %/V Test Conditions Min Typ Max Unit VC = 1.25 V IVC = 25 mA IVC = 5.0 mA (Note 3) (Note 3) FB = 1.0 V or NFB = -1.9 V, VC = 1.25 V FB = 1.5 V or NFB = -3.1 V, VC = 1.25 V FB = 1.0 V or NFB = -1.9 V; VC sources 25 mA FB = 1.5 V or NFB = -3.1 V, VC sinks 25 mA Reduce VC from 1.5 V until switching stops -0.05 300 115 200 100 -90 200 1.5 0.30 0.70 0.01 550 160 500 180 -50 460 1.64 0.47 1.05 0.05 800 225 - 320 -25 1500 1.9 0.70 1.30 %/V mMho mMho V/V V/V mA mA V V V Shutdown Threshold Shutdown Delay 2. For the FR4 suffix parts, production testing is performed at 25C and 85C; limits at 0C are guaranteed by design. 3. Guaranteed by design, not 100% tested in production. http://onsemi.com 3 NCP1442, NCP1443, NCP1444, NCP1445 ELECTRICAL CHARACTERISTICS (continued) (2.7 V < VCC < 30 V; 0C < TA < 85C; 0C < TJ < 125C; For all NCP1442/3/4/5 specifications unless otherwise stated.) (See Note 2) Characteristic Power Switch Switch Saturation Voltage ISWITCH = 4.0 A ISWITCH = 10 mA, 2.7 V < VCC < 12 V ISWITCH = 10 mA, 12 V < VCC < 30 V 50% duty cycle (Note 4) 80% duty cycle (Note 4) FB = 0 V or NFB = 0 V, ISW = 4.0 A (Note 4) 2.7 V VCC 12 V, 10 mA ISW 4.0 A 12 V < VCC 30 V, 10 mA ISW 4.0 A VSW = 40 V, VCC = 0V - - - 5.0 4.0 200 - - - 0.6 0.14 0.9 6.0 - 250 8.0 10 2.0 1.0 0.5 0.4 8.0 - 300 30 50 20 V V V A A ns mA/A mA Test Conditions Min Typ Max Unit Switch Current Limit Minimum Pulse Width Switch Transconductance, DICC/ DIVSW Switch Leakage General Operating Current Shutdown Mode Current ISW = 0 VC < 0.8 V, SS = 0 V, 2.7 V VCC 12 V VC < 0.8 V, SS = 0 V, 12 V VCC 30 V VSW switching, maximum ISW = 10 mA (Note 4) (Note 4) - - - - 150 - 15 16 25 2.2 180 25 27 60 60 2.6 210 - mA mA Minimum Operation Input Voltage Thermal Shutdown Thermal Hysteresis V C C 4. Guaranteed by design, not 100% tested in production. PACKAGE PIN DESCRIPTION Package Pin Number 1 Pin Symbol VC Function Loop compensation pin. The VC pin is the output of the error amplifier and is used for loop compensation, current limit and soft start. Loop compensation can be implemented by a simple RC network as shown in the application diagram on page 2. Positive regulator feedback pin. This pin senses a positive output voltage and is referenced to 1.276 V. When the voltage at this pin falls below 0.4 V, chip switching frequency reduces to 20% of the nominal frequency. These pins are connected to internal test logic and should either be left floating or tied to ground. Connection to a voltage between 2.0 V and 6.0 V shuts down the internal oscillator and leaves the power switch running. Negative feedback pin. This pin senses a negative output voltage and is referenced to -2.475 V. When the voltage at this pin goes above -0.65 V, chip switching frequency reduces to 20% of the nominal frequency. Ground pin. This pin provides a ground for the controller circuitry and the internal power switch. This pin is internally connected to the metal pad of the package to provide an additional ground connection as well as an effective means of dissipating heat. High current switch pin. This pin connects internally to the collector of the power switch. The open voltage across the power switch can be as high as 40 V. To minimize radiation, use a trace as short as practical. Synchronization and shutdown pin. This pin may be used to synchronize the part to nearly twice the base frequency. A TTL low will shut the part down and put it into low current mode. If synchronization is not used, this pin should be either tied high or left floating for normal operation. Input power supply pin. This pin supplies power to the part and should have a bypass capacitor connected to GND. 2 (NCP1442/4 only) FB 2 (NCP1443/5 only) 3 (NCP1442/4 only) 3 (NCP1443/5 only) Test NFB 4 GND 5 VSW 6 SS 7 VCC http://onsemi.com 4 NCP1442, NCP1443, NCP1444, NCP1445 VCC Shutdown Delay Timer SS Sync 2.0 V Regulator Thermal Shutdown VSW Oscillator S PWM Latch R Q Driver Switch Frequency Shift 5:1 x5 200 k 2.0 V Negative Error Amp + - Slope Compensation 15 mW Ramp Summer PWM Comparator + - NFB NCP1443/5 only 250 k GND -0.65 V Detector 0.4 V Detector FB NCP1442/4 only 1.276 V - + Positive Error Amp VC Figure 2. Block Diagram http://onsemi.com 5 NCP1442, NCP1443, NCP1444, NCP1445 20 VCC = 30 V ICC, SUPPLY CURRENT (mA) VCC = 12 V DICC / DISW (mA/A) 15 VCC = 2.7 V 10 10 8 6 VCC = 2.7 V 4 2 DISW = 2.99 A 0 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) 0 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) VCC = 30 V VCC = 12 V 12 5 Figure 3. Supply Current versus Temperature Figure 4. DICC / DISW versus Temperature VCE(sat), SWITCH SATURATION VOLTAGE (mV) 800 f, SWITCHING FREQUENCY (kHz) 700 600 500 400 300 200 100 0 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 ISW, SWITCH CURRENT (A) VCC = 2.7 V TA = 25C TA = 85C 300 295 290 285 280 275 270 265 260 255 250 0 10 20 30 40 50 60 70 80 TA, AMBIENT TEMPERATURE (C) Figure 5. Switch Saturation Voltage versus Switch Current Figure 6. Switching Frequency versus Temperature (NCP1442/3 Only) f, SWITCHING FREQUENCY (% of Typical) 600 f, SWITCHING FREQUENCY (kHz) 595 590 585 580 575 570 565 560 555 550 0 10 20 30 40 50 60 70 80 125 100 TA = 25C 75 TA = 85C 50 25 VCC = 12 V 0 0.38 0.39 0.40 0.41 0.42 0.43 0.44 0.45 VFB, POSITIVE FEEDBACK VOLTAGE (V) TA, AMBIENT TEMPERATURE (C) Figure 7. Switching Frequency versus Temperature (NCP1444/5 Only) Figure 8. Switching Frequency versus Positive Feedback Voltage http://onsemi.com 6 NCP1442, NCP1443, NCP1444, NCP1445 f, SWITCHING FREQUENCY (% of Typical) 125 1.276 1.275 1.274 1.273 VCC = 12 V VCC = 2.7 V 1.272 1.271 1.270 1.269 1.268 0 10 20 30 40 50 60 70 80 VCC = 30 V 100 TA = 85C 75 TA = 25C 50 25 0 -0.665 -0.67 VCC = 12 V -0.675 -0.68 -0.685 -0.69 -0.695 VFB, FEEDBACK REFERENCE VOLTAGE (V) VNFB, NEGATIVE FEEDBACK VOLTAGE (V) TA, AMBIENT TEMPERATURE (C) Figure 9. Switching Frequency versus Negative Feedback Voltage IFB, ERROR AMPLIFIER BIAS CURRENT (mA) VFB, FEEDBACK REFERENCE VOLTAGE (V) Figure 10. Feedback Reference Voltage versus Temperature (NCP1442/4 Only) 0.30 0.29 0.28 0.27 0.26 0.25 0.24 0.23 0.22 0.21 0.20 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) VCC = 30 V VCC = 2.7 V VCC = 12 V -2.46 -2.47 VCC = 12 V -2.48 VCC = 2.7 V -2.49 VCC = 30 V -2.50 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) Figure 11. Feedback Reference Voltage versus Temperature (NCP1443/5 Only) Figure 12. Error Amplifier Bias Current versus Temperature (NCP1442/3 Only) INFB, ERROR AMPLIFIER BIAS CURRENT (mA) -8 VCC = 30 V -9 12 V -10 2.7 V -11 -12 -13 -14 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) Dmax, MAXIMUM DUTY CYCLE (%) 97.0 96.5 96.0 95.5 95.0 94.5 94.0 93.5 93.0 0 20 40 VCC = 30 V 60 80 TA, AMBIENT TEMPERATURE (C) VCC = 2.7 V VCC = 12 V Figure 13. Error Amplifier Bias Current versus Temperature (NCP1443/5 Only) http://onsemi.com 7 Figure 14. Maximum Duty Cycle versus Temperature NCP1442, NCP1443, NCP1444, NCP1445 1.0 VSS, SHUTDOWN THRESHOLD (V) 0 10 20 30 40 50 60 70 80 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 0 10 20 30 40 50 60 70 80 1.14 Vcth, THRESHOLD VOLTAGE (V) 1.12 1.10 1.08 1.06 1.04 1.02 1.00 TA, AMBIENT TEMPERATURE (C) TA, AMBIENT TEMPERATURE (C) Figure 15. VC Threshold Voltage versus Temperature Figure 16. Shutdown Threshold versus Temperature 180 TD, SHUTDOWN DELAY (ms) TD, SHUTDOWN DELAY (ms) 160 140 120 100 80 60 40 20 0 0 10 20 30 40 50 60 70 80 TA, AMBIENT TEMPERATURE (C) VCC = 30 V VCC = 12 V VCC = 2.7 V 250 VCC = 2.7 V 200 150 100 VCC = 12 V 50 0 0 VCC = 30 V 10 20 30 40 50 60 70 80 TA, AMBIENT TEMPERATURE (C) Figure 17. Shutdown Delay versus Temperature (NCP1442) Figure 18. Shutdown Delay versus Temperature (NCP1444) 4.0 ISD, SUPPLY CURRENT (mA) 3.5 3.0 ISS (mA) 2.5 2.0 1.5 1.0 0.5 0 0 5 10 15 VSS (V) 20 25 30 45 40 35 30 25 20 15 10 5 0 0 5 10 15 20 25 30 VCC, SUPPLY VOLTAGE (V) TA = 25C TA = 85C Figure 19. ISS versus VSS Figure 20. Supply Current versus Supply Voltage During Shutdown http://onsemi.com 8 NCP1442, NCP1443, NCP1444, NCP1445 570 gm, TRANSCONDUCTANCE (mmho) 560 550 540 530 520 510 500 490 480 470 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) gm, TRANSCONDUCTANCE (mmho) -600 -650 -700 -750 -800 -850 -900 -950 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) Figure 21. Error Amplifier Transconductance versus Temperature Figure 22. Negative Error Amplifier Transconductance versus Temperature 100 IC, EA OUTPUT CURRENT (mA) 0 -100 -200 -300 -400 -500 -0.25 -0.2 -0.15 -0.1 -0.05 IC, EA OUTPUT CURRENT (mA) 0.05 0.1 0.15 0.2 0.25 100 50 0 -50 -100 -150 -200 -0.3 0 -0.2 -0.1 0 0.1 0.2 Vref-VFB, FEEDBACK VOLTAGE (mV) Vref-VNFB, FEEDBACK VOLTAGE (mV) Figure 23. Error Amplifier Output Current versus Positive Feedback Voltage Figure 24. Error Amplifier Output Current versus Negative Feedback Voltage ISW, SWITCH LEAKAGE CURRENT (mA) 6.0 5.5 5.0 4.5 4.0 3.5 3.0 0 20 40 60 80 TA, AMBIENT TEMPERATURE (C) Figure 25. Switch Leakage Current versus Temperature http://onsemi.com 9 NCP1442, NCP1443, NCP1444, NCP1445 APPLICATIONS INFORMATION THEORY OF OPERATION Current Mode Control VCC Oscillator S VC - + PWM Comparator Q Power Switch L R D1 VSW In Out X5 Driver CO RLOAD SUMMER Slope Compensation 15 mW Figure 26. Current Mode Control Scheme The NCP144X family incorporates a current mode control scheme, in which the PWM ramp signal is derived from the power switch current. This ramp signal is compared to the output of the error amplifier to control the on-time of the power switch. The oscillator is used as a fixed-frequency clock to ensure a constant operational frequency. The resulting control scheme features several advantages over conventional voltage mode control. First, derived directly from the inductor, the ramp signal responds immediately to line voltage changes. This eliminates the delay caused by the output filter and error amplifier, which is commonly found in voltage mode controllers. The second benefit comes from inherent pulse-by-pulse current limiting by merely clamping the peak switching current. Finally, since current mode commands an output current rather than voltage, the filter offers only a single pole to the feedback loop. This allows both a simpler compensation and a higher gain-bandwidth over a comparable voltage mode circuit. Without discrediting its apparent merits, current mode control comes with its own peculiar problems, mainly, subharmonic oscillation at duty cycles over 50%. The NCP144X family solves this problem by adopting a slope compensation scheme in which a fixed ramp generated by the oscillator is added to the current ramp. A proper slope rate is provided to improve circuit stability without sacrificing the advantages of current mode control. Oscillator and Shutdown Sync Current Ramp VSW The oscillator is trimmed to guarantee frequency accuracy. The output of the oscillator turns on the power switch at a frequency of 280 kHz (NCP1442/3) or 560 kHz (NCP1444/5), as shown in Figure 26. The power switch is turned off by the output of the PWM Comparator. A TTL-compatible sync input at the SS pin is capable of syncing up to 1.8 times the base oscillator frequency. As shown in Figure 27, in order to sync to a higher frequency, a positive transition turns on the power switch before the output of the oscillator goes high, thereby resetting the oscillator. The sync operation allows multiple power supplies to operate at the same frequency. A sustained logic low at the SS pin will shut down the IC and reduce the supply current. An additional feature includes frequency shift to 20% of the nominal frequency when either the NFB or FB pins trigger the threshold. During power up, overload, or short circuit conditions, the minimum switch on-time is limited by the PWM comparator minimum pulse width. Extra switch off-time reduces the minimum duty cycle to protect external components and the IC itself. As previously mentioned, this block also produces a ramp for the slope compensation to improve regulator stability. Error Amplifier 200 k NFB 250 k + NCP1443/5 - 2.0 V negative error-amp 1MW 120 pF VC FB 1.276 V + - NCP1442/4 Voltage Clamp C1 0.01 mF R1 5 kW positive error-amp Figure 28. Error Amplifier Equivalent Circuit Figure 27. Timing Diagram of Sync and Shutdown For NCP1443/5, the NFB pin is internally referenced to -2.475 V with approximately a 250 kW input impedance. For NCP1442/4, the FB pin is directly connected to the inverting input of the positive error amplifier, whose non-inverting input is fed by the 1.276 V reference. Both amplifiers are transconductance amplifiers with a high output impedance of approximately 1.0 MW, as shown in Figure 28. The VC pin is connected to the output of the error amplifiers and is internally clamped between 0.5 V and 1.7 V. A typical connection at the VC pin includes a capacitor in series with a resistor to ground, forming a pole/zero for loop compensation. An external shunt can be connected between the VC pin and ground to reduce its clamp voltage. Consequently, the current limit of the internal power transistor current is reduced from its nominal value. http://onsemi.com 10 NCP1442, NCP1443, NCP1444, NCP1445 Switch Driver and Power Switch The switch driver receives a control signal from the logic section to drive the output power switch. The switch is grounded through emitter resistors (15 mW total) to the GND pin. The peak switching current is clamped by an internal circuit. The clamp current is guaranteed to be greater than 4.0 A and varies with duty cycle due to slope compensation. The power switch can withstand a maximum voltage of 40 V on the collector (VSW pin). The saturation voltage of the switch is typically less than 1.0 V to minimize power dissipation. Short Circuit Condition When a short circuit condition happens in a boost circuit, the inductor current will increase during the whole switching cycle, causing excessive current to be drawn from the input power supply. Since control ICs don't have the means to limit load current, an external current limit circuit (such as a fuse or relay) has to be implemented to protect the load, power supply and ICs. In other topologies, the frequency shift built into the IC prevents damage to the chip and external components. This feature reduces the minimum duty cycle and allows the transformer secondary to absorb excess energy before the switch turns back on. output through the inductor and diode. Once VCC reaches approximately 1.5 V, the internal power switch briefly turns on. This is a part of the NCP144X's normal operation. The turn-on of the power switch accounts for the initial current swing. When the VC pin voltage rises above the threshold, the internal power switch starts to switch and a voltage pulse can be seen at the VSW pin. Detecting a low output voltage at the FB pin, the built-in frequency shift feature reduces the switching frequency to a fraction of its nominal value, reducing the minimum duty cycle, which is otherwise limited by the minimum on-time of the switch. The peak current during this phase is clamped by the internal current limit. When the FB pin voltage rises above 0.4 V, the frequency increases to its nominal value, and the peak current begins to decrease as the output approaches the regulation voltage. The overshoot of the output voltage is prevented by the active pull-on, by which the sink current of the error amplifier is increased once an overvoltage condition is detected. The overvoltage condition is defined as when the FB pin voltage is 50 mV greater than the reference voltage. COMPONENT SELECTION Frequency Compensation IL The goal of frequency compensation is to achieve desirable transient response and DC regulation while ensuring the stability of the system. A typical compensation network, as shown in Figure 30, provides a frequency response of two poles and one zero. This frequency response is further illustrated in the Bode plot shown in Figure 31. VOUT VCC VC VC NCP1442/3/4/5 C1 GND R1 C2 Figure 29. Startup Waveforms of Circuit Shown in the Application Diagram. Load = 400 mA. Figure 30. A Typical Compensation Network The NCP144X can be activated by either connecting the VCC pin to a voltage source or by enabling the SS pin. Startup waveforms shown in Figure 29 are measured in the boost converter demonstrated in the Block Diagram (Figure 2). Recorded after the input voltage is turned on, this waveform shows the various phases during the power up transition. When the VCC voltage is below the minimum supply voltage, the VSW pin is in high impedance. Therefore, current conducts directly from the input power source to the The high DC gain in Figure 31 is desirable for achieving DC accuracy over line and load variations. The DC gain of a transconductance error amplifier can be calculated as follows: GainDC + GM RO where: GM = error amplifier transconductance; RO = error amplifier output resistance 1.0 MW. http://onsemi.com 11 NCP1442, NCP1443, NCP1444, NCP1445 The low frequency pole, fP1, is determined by the error amplifier output resistance and C1 as: 1 fP1 + 2pC1R O -VOUT 2V RP 200 kW NFB R2 RIN 250 kW R1 The first zero generated by C1 and R1 is: 1 fZ1 + 2pC1R1 + - Negative Error-Amp The phase lead provided by this zero ensures that the loop has at least a 45 phase margin at the crossover frequency. Therefore, this zero should be placed close to the pole generated in the power stage which can be identified at frequency: 1 fP + 2pCORLOAD Figure 32. Negative Error Amplifier and NFB Pin where: CO = equivalent output capacitance of the error amplifier 120pF; RLOAD= load resistance. The high frequency pole, fP2, can be placed at the output filter's ESR zero or at half the switching frequency. Placing the pole at this frequency will cut down on switching noise. The frequency of this pole is determined by the value of C2 and R1: 1 fP2 + 2pC2R1 It is shown that if R1 is less than 10 k, the deviation from the design target will be less than 0.1 V. If the tolerances of the negative voltage reference and NFB pin input current are considered, the possible offset of the output VOFFSET varies in the range of: *0.0.5 (R1 ) R2) * (15 mA R1) v VOFFSET R2 0.0.5 (R1 ) R2) * (5 mA R1) v R2 VSW Voltage Limit One simple method to ensure adequate phase margin is to design the frequency response with a -20 dB per decade slope, until unity-gain crossover. The crossover frequency should be selected at the midpoint between fZ1 and fP2 where the phase margin is maximized. DC Gain In the boost topology, VSW pin maximum voltage is set by the maximum output voltage plus the output diode forward voltage. The diode forward voltage is typically 0.5 V for Schottky diodes and 0.8 V for ultrafast recovery diodes: VSW(MAX) + VOUT(MAX))VF where: VF = output diode forward voltage. In the flyback topology, peak VSW voltage is governed by: VSW(MAX) + VCC(MAX))(VOUT)VF) N fP1 fZ1 fP2 Frequency (LOG) Figure 31. Bode Plot of the Compensation Network Shown in Figure 30 Negative Voltage Feedback where: N = transformer turns ratio, primary over secondary. When the power switch turns off, there exists a voltage spike superimposed on top of the steady-state voltage. Usually this voltage spike is caused by transformer leakage inductance charging stray capacitance between the VSW and GND pins. To prevent the voltage at the VSW pin from exceeding the maximum rating, a transient voltage suppressor in series with a diode is paralleled with the primary windings. Another method of clamping switch voltage is to connect a transient voltage suppressor between the VSW pin and ground. Since the negative error amplifier has finite input impedance as shown in Figure 32, its induced error has to be considered. If a voltage divider is used to scale down the negative output voltage for the NFB pin, the equation for calculating output voltage is: *VOUT + *2.475 (R1 ) R2) *10 mA R2 R1 Gain (dB) http://onsemi.com 12 NCP1442, NCP1443, NCP1444, NCP1445 Magnetic Component Selection IIN IL When choosing a magnetic component, one must consider factors such as peak current, core and ferrite material, output voltage ripple, EMI, temperature range, physical size and cost. In boost circuits, the average inductor current is the product of output current and voltage gain (VOUT/VCC), assuming 100% energy transfer efficiency. In continuous conduction mode, inductor ripple current is: V (V * VCC) IRIPPLE + CC OUT (f)(L)(VOUT) VCC + - CIN RESR where: f = 280 kHz for NCP1442/3 and 560 kHz for NCP1444/5. The peak inductor current is equal to average current plus half of the ripple current, which should not cause inductor saturation. The above equation can also be referenced when selecting the value of the inductor based on the tolerance of the ripple current in the circuits. Small ripple current provides the benefits of small input capacitors and greater output current capability. A core geometry like a rod or barrel is prone to generating high magnetic field radiation, but is relatively cheap and small. Other core geometries, such as toroids, provide a closed magnetic loop to prevent EMI. Input Capacitor Selection Figure 34. Boost Circuit Effective Input Filter The situation is different in a flyback circuit. The input current is discontinuous and a significant pulsed current is seen by the input capacitors. Therefore, there are two requirements for capacitors in a flyback regulator: energy storage and filtering. To maintain a stable voltage supply to the chip, a storage capacitor larger than 20 mF with low ESR is required. To reduce the noise generated by the inductor, insert a 1.0 mF ceramic capacitor between VCC and ground as close as possible to the chip. Output Capacitor Selection In boost circuits, the inductor becomes part of the input filter, as shown in Figure 34. In continuous mode, the input current waveform is triangular and does not contain a large pulsed current, as shown in Figure 33. This reduces the requirements imposed on the input capacitor selection. During continuous conduction mode, the peak to peak inductor ripple current is given in the previous section. As we can see from Figure 33, the product of the inductor current ripple and the input capacitor's effective series resistance (ESR) determine the VCC ripple. In most applications, input capacitors in the range of 10 mF to 100 mF with an ESR less than 0.3 W work well up to a full 4.0 A switch current. VOUT ripple IL Figure 35. Typical Output Voltage Ripple VCC ripple IIN IL Figure 33. Boost Input Voltage and Current Ripple Waveforms By examining the waveforms shown in Figure 35, we can see that the output voltage ripple comes from two major sources, namely capacitor ESR and the charging/discharging of the output capacitor. In boost circuits, when the power switch turns off, IL flows into the output capacitor causing an instant DV = IIN x ESR. At the same time, current IL - IOUT charges the capacitor and increases the output voltage gradually. When the power switch is turned on, IL is shunted to ground and IOUT discharges the output capacitor. When the IL ripple is small enough, IL can be treated as a constant and is equal to input http://onsemi.com 13 NCP1442, NCP1443, NCP1444, NCP1445 current IIN. Summing up, the output voltage peak-peak ripple can be calculated by: (I * IOUT)(1 * D) VOUT(RIPPLE) + IN (COUT)(f) I D ) OUT ) IIN ESR (COUT)(f) VIN VCC R2 VC D1 The equation can be expressed more conveniently in terms of VCC, VOUT and IOUT for design purposes as follows: I (V * VCC) VOUT(RIPPLE) + OUT OUT (COUT)(f) (I )(V )(ESR) ) OUT OUT VCC 1 (COUT)(f) R3 R1 C1 C2 The capacitor RMS ripple current is: IRIPPLE + (IIN * IOUT)2(1 * D))(IOUT)2(D) + IOUT VOUT * VCC VCC Figure 36. Current Limiting using a Diode Clamp Although the above equations apply only for boost circuits, similar equations can be derived for flyback circuits. Reducing the Current Limit Another solution to the current limiting problem is to externally measure the current through the switch using a sense resistor. Such a circuit is illustrated in Figure 37. VCC VIN VC + ISWREAV where: RE = .015 W, the value of the internal emitter resistor; AV = 5.0 V/V, the gain of the current sense amplifier. Since RE and AV cannot be changed by the end user, the only available method for limiting switch current below 4.0 A is to clamp the VC pin at a lower voltage. If the maximum switch or inductor current is substituted into the equation above, the desired clamp voltage will result. A simple diode clamp, as shown in Figure 36, clamps the VC voltage to a diode drop above the voltage on resistor R3. Unfortunately, such a simple circuit is not generally acceptable if VIN is loosely regulated. Figure 37. Current Limiting using a Current Sense Resistor The switch current is limited to: VBE(Q1) ISWITCH(PEAK) + RSENSE where: VBE(Q1) = the base-emitter voltage drop of Q1, typically 0.65 V. http://onsemi.com 14 - + In some applications, the designer may prefer a lower limit on the switch current than 4.0 A. An external shunt can be connected between the VC pin and ground to reduce its clamp voltage. Consequently, the current limit of the internal power transistor current is reduced from its nominal value. The voltage on the VC pin can be evaluated with the equation: PGND AGND VC R1 Q1 R2 C3 RSENSE Output Ground C1 C2 NCP1442, NCP1443, NCP1444, NCP1445 The improved circuit does not require a regulated voltage to operate properly. Unfortunately, a price must be paid for this convenience in the overall efficiency of the circuit. The designer should note that the input and output grounds are no longer common. Also, the addition of the current sense resistor, RSENSE, results in a considerable power loss which increases with the duty cycle. Resistor R2 and capacitor C3 form a low-pass filter to remove noise. Subharmonic Oscillation Subharmonic oscillation (SHM) is a problem found in current-mode control systems, where instability results when duty cycle exceeds 50%. SHM only occurs in switching regulators with a continuous inductor current. This instability is not harmful to the converter and usually does not affect the output voltage regulation. SHM will increase the radiated EM noise from the converter and can cause, under certain circumstances, the inductor to emit high-frequency audible noise. SHM is an easily remedied problem. The rising slope of the inductor current is supplemented with internal "slope compensation" to prevent any duty cycle instability from carrying through to the next switching cycle. In the NCP144X family, slope compensation is added during the entire switch on-time, typically in the amount of 180 mA/ms. In some cases, SHM can rear its ugly head despite the presence of the onboard slope compensation. The simple cure to this problem is more slope compensation to avoid the unwanted oscillation. In that case, an external circuit, shown in Figure 38, can be added to increase the amount of slope compensation used. This circuit requires only a few components and is "tacked on" to the compensation network. VSW VSW The dashed box contains the normal compensation circuitry to limit the bandwidth of the error amplifier. Resistors R2 and R3 form a voltage divider off of the VSW pin. In normal operation, VSW looks similar to a square wave, and is dependent on the converter topology. Formulas for calculating VSW in the boost and flyback topologies are given in the section "VSW Voltage Limit." The voltage on VSW charges capacitor C3 when the switch is off, causing the voltage at the VC pin to shift upwards. When the switch turns on, C3 discharges through R3, producing a negative slope at the VC pin. This negative slope provides the slope compensation. The amount of slope compensation added by this circuit is R3 DI + V SW R )R DT 2 3 *(1*D) 1 * e R3C3fSW fSW (1 * D)REAV where: DI/DT = the amount of slope compensation added (A/s); VSW = the voltage at the switch node when the transistor is turned off (V); fSW = the switching frequency, typically 280 kHz (NCP1442/3) or 560 kHz (NCP1444/5) (Hz); D = the duty cycle; RE = 0.015 W, the value of the internal emitter resistor; AV = 5.0 V/V, the gain of the current sense amplifier. In selecting appropriate values for the slope compensation network, the designer is advised to choose a convenient capacitor, then select values for R2 and R3 such that the amount of slope compensation added is 100 mA/ms. Then R2 may be increased or decreased as necessary. Of course, the series combination of R2 and R3 should be large enough to avoid drawing excessive current from VSW. Additionally, to ensure that the control loop stability is improved, the time constant formed by the additional components should be chosen such that: R3C3 t 1 * D fSW VC R1 C1 C2 R2 Finally, it is worth mentioning that the added slope compensation is a trade-off between duty cycle stability and transient response. The more slope compensation a designer adds, the slower the transient response will be, due to the external circuitry interfering with the proper operation of the error amplifier. Soft-Start C3 R3 Figure 38. Technique for Increasing Slope Compensation Through the addition of an external circuit, a soft-start function can be added to the NCP1442/3/4/5 family of components. Soft-start circuitry prevents the VC pin from slamming high during startup, thereby inhibiting the inductor current from rising at a high slope. This circuit, shown in Figure 39, requires a minimum number of components and allows the soft-start circuitry to activate any time the SS pin is used to restart the converter. http://onsemi.com 15 NCP1442, NCP1443, NCP1444, NCP1445 VIN VCC SS SS VC The internal control circuitry, including the oscillator and linear regulator, requires a small amount of power even when the switch is turned off. The specifications section of this datasheet reveals that the typical operating current, IQ, due to this circuitry is 5.5 mA. Additional guidance can be found in the graph of operating current vs. temperature. This graph shows that IQ is strongly dependent on input voltage, VIN, and the ambient temperature, TA. Then: PBIAS + VINIQ R1 C1 D1 D2 C3 C2 Since the onboard switch is an NPN transistor, the base drive current must be factored in as well. This current is drawn from the VIN pin, in addition to the control circuitry current. The base drive current is listed in the specifications as DICC/DISW, or switch transconductance. As before, the designer will find additional guidance in the graphs. With that information, the designer can calculate: PDRIVER + VINISW ICC DISW D Figure 39. Soft-Start Resistor R1 and capacitors C1 and C2 form the compensation network. At turn on, the voltage at the VC pin starts to come up, charging capacitor C3 through Schottky diode D2, clamping the voltage at the VC pin such that switching begins when VC reaches the VC threshold, typically 1.05 V (refer to graphs for detail over temperature). VC + VF(D2))VC3 where: ISW = the current through the switch; D = the duty cycle or percentage of switch on-time. ISW and D are dependent on the type of converter. In a boost converter, ISW(AVG) ^ ILOAD D I efficiency V * VIN D ^ OUT VOUT Therefore, C3 slows the startup of the circuit by limiting the voltage on the VC pin. The soft-start time increases with the size of C3. Diode D1 discharges C3 when SS is low. If the shutdown function is not used with this part, the cathode of D1 should be connected to VIN. Calculating Junction Temperature In a flyback converter, V I ISW(AVG) ^ OUT LOAD VIN D^ VOUT ns VOUT ) np VIN I efficiency To ensure safe operation of the NCP1442/3/4/5, the designer must calculate the on-chip power dissipation and determine its expected junction temperature. Internal thermal protection circuitry will turn the part off once the junction temperature exceeds 180C 30. However, repeated operation at such high temperatures will ensure a reduced operating life. Calculation of the junction temperature is an imprecise but simple task. First, the power losses must be quantified. There are three major sources of power loss on the NCP144X: * biasing of internal control circuitry, PBIAS * switch driver, PDRIVER * switch saturation, PSAT where: ns = number of turns in the transformer secondary winding. np = number of turns in the transformer primary winding. The switch saturation voltage, V(CE)SAT, is the last major source of on-chip power loss. V(CE)SAT is the collector-emitter voltage of the internal NPN transistor when it is driven into saturation by its base drive current. The value for V(CE)SAT can be obtained from the specifications or from the graphs, as "Switch Saturation Voltage." Thus, PSAT ^ V(CE)SATISW D Finally, the total on-chip power losses are: PD + PBIAS)PDRIVER)PSAT http://onsemi.com 16 NCP1442, NCP1443, NCP1444, NCP1445 Power dissipation in a semiconductor device results in the generation of heat in the junctions at the surface of the chip. This heat is transferred to the surface of the IC package, but a thermal gradient exists due to the resistive properties of the package molding compound. The magnitude of the thermal gradient is expressed in manufacturers' data sheets as qJA, or junction-to-ambient thermal resistance. The on-chip junction temperature can be calculated if qJA, the air temperature near the surface of the IC, and the on-chip power dissipation are known. TJ + TA)(PDqJA) surface of the chip might be considered to reduce TA. A copper "landing pad" can be connected to ground - designers are referred to ON Semiconductor applications note SR006 for more information on properly sizing a copper area. Circuit Layout Guidelines In any switching power supply, circuit layout is very important for proper operation. Rapidly switching currents combined with trace inductance generates voltage transitions that can cause problems. Therefore the following guidelines should be followed in the layout. 1. In boost circuits, high AC current circulates within the loop composed of the diode, output capacitor, and on-chip power transistor. The length of associated traces and leads should be kept as short as possible. In the flyback circuit, high AC current loops exist on both sides of the transformer. On the primary side, the loop consists of the input capacitor, transformer, and on-chip power transistor, while the transformer, rectifier diodes, and output capacitors form another loop on the secondary side. Just as in the boost circuit, all traces and leads containing large AC currents should be kept short. 2. Separate the low current signal grounds from the power grounds. Use single point grounding or ground plane construction for the best results. 3. Locate the voltage feedback resistors as near the IC as possible to keep the sensitive feedback wiring short. Connect feedback resistors to the low current analog ground. where: TJ = IC or FET junction temperature (C); TA = ambient temperature (C); PD = power dissipated by part in question (W); qJA = junction-to-ambient thermal resistance (C/W). For ON Semiconductor components, the value for qJA can be found on page 19 of the datasheet, under "Package Thermal Data." Note that this value is different for every package style and every manufacturer. For the NCP144X, qJA varies between 10-50C/W, depending upon the size of the copper pad to which the IC is mounted. Once the designer has calculated TJ, the question of whether the NCP144X can be used in an application is settled. If TJ exceeds 150C, the absolute maximum allowable junction temperature, the NCP144X is not suitable for that application. If TJ approaches 150C, the designer should consider possible means of reducing the junction temperature. Perhaps another converter topology could be selected to reduce the switch current. Increasing the airflow across the http://onsemi.com 17 NCP1442, NCP1443, NCP1444, NCP1445 ORDERING INFORMATION Device NCP1442FR4 NCP1442FR4G NCP1442T NCP1443FR4 NCP1443FR4G 0C < TA < 85C NCP1443T NCP1444FR4 NCP1444T NCP1445FR4 NCP1445T Operating Temperature Range Package 7 Lead PowerFLEX Short-Leaded 7 Lead PowerFLEX Short-Leaded (Pb-Free) 7 Lead TO-220 (Straight Lead) 7 Lead PowerFLEX Short-Leaded 7 Lead PowerFLEX Short-Leaded (Pb-Free) 7 Lead TO-220 (Straight Lead) 7 Lead PowerFLEX Short-Leaded 7 Lead TO-220 (Straight Lead) 7 Lead PowerFLEX Short-Leaded 7 Lead TO-220 (Straight Lead) Shipping 2000 Tape & Reel 2000 Tape & Reel 50 Units/Rail 2000 Tape & Reel 2000 Tape & Reel 50 Units/Rail 2000 Tape & Reel 50 Units/Rail 2000 Tape & Reel 50 Units/Rail For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D. http://onsemi.com 18 NCP1442, NCP1443, NCP1444, NCP1445 PACKAGE DIMENSIONS PowerFLEX 7-PIN F SUFFIX CASE 936J-01 ISSUE O A AE J B PN C L M NOTES: 1. DIMENSIONS AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSIONS A AND B DO NOT INCLUDE MOLD FLASH OR GATE PROTRUSIONS. MOLD FLASH AND GATE PROTRUSIONS NOT TO EXCEED 0.025 (0.635) MAX. INCHES MIN MAX 0.350 0.360 0.350 0.360 0.070 0.080 0.026 0.030 0.005 0.015 0.031 0.041 0.050 BSC 0.008 0.012 0.410 0.420 0.365 00.375 0.040 REF 0.361 0.367 0.310 0.320 0.394 0.400 0.002 --- 0.070 0.080 0.001 0.005 12 0.296 REF 0.075 REF 0.071 REF 0.140 REF 0.220 REF 0.281 REF 12 3 6 MILLIMETERS MIN MAX 8.89 9.14 8.89 9.14 1.78 2.03 0.66 0.76 0.13 0.38 0.79 1.04 1.270 BSC 0.199 0.301 10.41 10.67 9.27 9.53 1.02 REF 9.16 9.31 7.87 8.13 10.00 10.16 0.05 --- 1.78 2.03 0.03 0.13 12 7.52 REF 1.91 REF 1.81 REF 3.56 REF 5.58 REF 7.14 REF 12 3 6 D 7 PL G 7 PL DETAIL AG 0.076 (0.003) S K -T- SEATING PLANE R 0.25 (0.010) R (TOP OFFSET) V H THERMAL DIE PAD E U AF F R 0.20 (0.008) W Y AB AC AD AA DETAIL AG DIM A B C D E F G H J K L M N P R S U V W Y AA AB AC AD AE AF PACKAGE THERMAL DATA Parameter RqJC RqJA Typical Typical PowerFLEX 7-PIN 1.0-4.0 10-50* Unit C/W C/W *Depending on thermal properties of substrate. RqJA = RqJC + RqCA. http://onsemi.com 19 NCP1442, NCP1443, NCP1444, NCP1445 PACKAGE DIMENSIONS 7 LEAD TO-220 T SUFFIX CASE 821P-03 ISSUE B C E N Q U L K A M NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. 821P-01 AND -02 OBSOLETE. NEW STANDARD IS 821P-03. DIM A B C D E F G H J K L M N Q U MILLIMETERS MIN MAX 9.91 10.54 8.23 9.40 4.19 4.83 0.66 0.81 0.89 1.40 7.62 TYP 1.22 1.32 2.16 2.92 0.30 0.64 24.00 26.54 26.67 29.03 6.10 6.48 7 --- 3.53 3.96 4 6 INCHES MIN MAX 0.390 0.415 0.324 0.370 0.165 0.190 0.026 0.032 0.035 0.055 0.3 TYP 0.048 0.052 0.085 0.115 0.012 0.025 0.945 1.045 1.050 1.143 0.240 0.255 7 --- 0.139 0.156 4 6 B J H D 7 PL G F 6 PL PowerFLEX is a trademark of Texas Instruments Incorporated. ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. "Typical" parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner. PUBLICATION ORDERING INFORMATION LITERATURE FULFILLMENT: Literature Distribution Center for ON Semiconductor P.O. Box 61312, Phoenix, Arizona 85082-1312 USA Phone: 480-829-7710 or 800-344-3860 Toll Free USA/Canada Fax: 480-829-7709 or 800-344-3867 Toll Free USA/Canada Email: orderlit@onsemi.com N. American Technical Support: 800-282-9855 Toll Free USA/Canada Japan: ON Semiconductor, Japan Customer Focus Center 2-9-1 Kamimeguro, Meguro-ku, Tokyo, Japan 153-0051 Phone: 81-3-5773-3850 ON Semiconductor Website: http://onsemi.com Order Literature: http://www.onsemi.com/litorder For additional information, please contact your local Sales Representative. http://onsemi.com 20 NCP1442/D |
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